Asymmetric digital subscriber line (ADSL) systems are used to implement broadcast digital TV, on-demand video, high-speed video-based internet access, and other forms of data transfers over existing twisted-pair telephone lines. Recently, the International Telecommunication Union published a draft recommendation for an ADSL system. See "Splitterless Asymmetric Digital Subscriber Line (ADSL) Transceivers," International Telecommunication Union, ITU-T, G.992.2, March, 1999, which is incorporated herein by reference.
To provide reliable data transfers, ADSL uses DMT technology. Methods for encoding and decoding data using DMT technology in an ADSL system are well known. See, for example, J. Gibson, The Communications Handbook, pages 450-479, IEEE Press, 1997, which is incorporated herein by reference.
Generally, a DMT system uses a number of frequencies (or carriers) to transmit data over a communication channel. A block diagram of a DMT system is shown in FIG. 1. The DMT system includes encoder 100, communication channel 110 and decoder 120. Encoder 100 receives serial data through line 102. Encoder 100 encodes the data using a DMT system. Encoder 100 then provides the encoded data through line 104 to channel 110. Typically, channel 110 includes a telephone line. After transmission through channel 110, the encoded data is provided through line 112 to decoder 120. Decoder 120 attempts to create a replica of the original serial data received by the encoder 100 through line 102. To do so, decoder 120 attempts to minimize the effects of channel 110 on the encoded data. Decoder 120 also provides the complementary coding functionality to that of encoder 100.
In many applications, a transceiver connects to each end of channel 110. The transceiver includes both an encoder and a decoder. A portion of the frequency spectrum is allocated to each transceiver. This configuration allows data transmission in both directions.
Turning to FIG. 2, encoder 100 is described in further detail. The encoder 100 includes a serial-to-parallel converter 202. Serial-to-parallel converter 202 receives the serial data through line 102 at a rate of B bits/second. Serial-to-parallel converter 202 groups the serial data into blocks of complex data having a length of N. In a typical implementation, N is 256.
Channel characteristics will vary depending upon the specific implementation. For example, two different twisted-pair telephone connections will have, in all likelihood, different characteristics. Accordingly, encoder 100 determines the frequency characteristics of the channel and attempts to determine an optimal loading distribution. One method of determining a loading distribution is described in U.S. Pat. No. 5,479,447, "Method and Apparatus for Adaptive, Variable Bandwidth, High-Speed Data Transmission of a Multicarrier Signal over Digital Subscriber Lines," P. Chow et al., Dec. 26, 1995, which is incorporated herein by reference. Serial-to-parallel converter 202 distributes the serial data according to the loading distribution. The resulting parallel data, X.sub.0,k, X.sub.1,k, . . . X.sub.N-1,k (where k denotes the block number), is provided over lines 204 to inverse fast Fourier transform (IFFT) 206.
The IFFT 206 transforms the parallel data from the frequency domain into the time domain. As shown, IFFT 206 is 2N long. This transformation distributes the information contained in each element of the parallel data across the transformed data, X.sub.0,k, X.sub.1,k, . . . X.sub.2N-1,k.
The transformed data, X.sub.0,k, X.sub.1,k, . . . X.sub.2N-1,k, is provided over lines 208 to cyclic prefix adder 210. Cyclic prefix adder 210 attaches a cyclic prefix, X.sub.2N-1,k, X.sub.2N-2,k, . . . X.sub.2N-v,k, to each block of data. The FFT sub-channel outputs do not interfere with one another if the cyclic prefix is longer in length than the impulse response of the channel. In practical applications, the impulse response of the channel (e.g., a twisted pair) is infinite in length. As will be described, further below, time-domain equalization (TEQ) is applied by a decoder to reduce the effects of the channel to a substantially finite duration. The cyclic prefix, X.sub.2N-1,k, X.sub.2N-2,k, . . . X.sub.2N-v,k, is chosen so that it exceeds this finite duration. Together the TEQ and the cyclic prefix operated to minimize the effect of the channel on the transmitted data.
Cyclic prefix adder 210 provides the prefixed and transformed data, X.sub.2N-v,k, . . . X.sub.2N-2,k, X.sub.2N-1,k, X.sub.0,k, X.sub.1,k, . . . X.sub.2N-1,k, through lines 212 to parallel-to-serial converter 214. Parallel-to-serial converter 214 converts this data to a serial format and provides the resulting bit stream, x(t), through line 216 to digital-to-analog converter 218. Digital-to-analog converter 218 provides the resulting analog signal through line 220 to low-pass filter 222. Low-pass filter 222 removes high-frequency noise resulting from the conversion and provides the resulting signal through line 226 to isolation transformer 226. The output of isolation transformer 226 connects to the communication channel 110 (shown in FIG. 1) through line 104.
Turning to FIG. 3, a block diagram showing a model of communication channel 110 is described. Communication channel 110 is a digital subscriber line (DSL) or like medium suitable for exchanging data. As shown, communication channel 110 is modeled as a filter having an impulse response, h(t), with an additive noise component, n(t). According to this model, filter 302 convolves x(t) with h(t). In the frequency domain, h(t) corresponds to H(.omega.). The result is provided to adder 306. Adder 306 also receives noise component, n(t), from channel noise generator 304. Therefore, passing x(t) through communication channel 110 produces x(t)*h(t)+n(t), where "*" denotes the operation of convolution. The resulting signal, y(t), is provided to decoder 120 (shown in FIG. 1) through line 112.
Turning to FIG. 4, decoder 120 is shown in further detail. The received signal, y(t), is passed though transformer 410, which acts to isolate any D.C. components. The received signal is then provided through line 412 to low-pass filter 414. Low-pass filter 414 removes high-frequency noise. The signal is then provided through line 416 to analog-to-digital converter 418 where it is converted to a digital signal. The received signal is then provided through line 420 to time-domain equalizer 422. Time-domain equalizer 412 includes a finite-impulse-response (FIR) filter that limits the effective memory of channel 110. The signal is then passed over line 424 to serial-to-parallel converter 426. Serial-to-parallel converter passes the resulting parallel data, y.sub.2N-v,k, . . . y.sub.2N-2,k, y.sub.2N-1,k, y.sub.0,k, y.sub.1,k, . . . y.sub.2N-1,k, over lines 428 to cyclic prefix stripper 430. The resulting signal, y.sub.0,k, y.sub.1,k, y.sub.2N-1,k, is provided over lines 432 to fast Fourier transform (FFT) 434. The signal provided to FFT 144 is 2N wide and is transformed from a time domain representation to a frequency domain representation. Since the amplitude vs. frequency and the delay vs. frequency responses of the channel are not necessarily constant across the frequency band, the received signal, Y.sub.0,k, Y.sub.1,k, . . . Y.sub.N-1,k, will differ from the encoded signal X.sub.0,k, X.sub.1,k, . . . X.sub.N-1,k. Frequency domain equalizer (FEQ) 438 provides a simple form of compensation for these differences. The FEQ 438 individually adjusts the attenuation and delay of each of the carriers.
The FEQ 438 passes the resulting signals, Z.sub.0,k, Z.sub.1,k, . . . Z.sub.2N-1,k, through lines 440 to parallel-to-serial converter 442. Parallel-to-serial converter passes the resulting signal out through line 122. If the encoding, transmission, and decoding processes work properly, the signal passed through line 122 will match the signal received through line 102 (shown in FIG. 1).
As mentioned above, the channel 110 often includes a twisted-pair telephone connection. Due to the dispersive nature of such channels, severe channel attenuation as well as intersymbol interference (ISI) are unavoidable. The TEQ and FEQ coefficients are used to mitigate the effects of channel attenuation and ISI. Since the channel characteristics vary, the TEQ and the FEQ must be optimized on a channel-by-channel basis.
Various methods for generating the TEQ and FEQ coefficients are known in the art. For example, U.S. Pat. No. 5,285,474, "Method for Equalizing a Multicarrier Signal in a Multicarrier Communication System," J. Chow et al., Feb. 8, 1994, which is incorporated herein by reference, teaches a method for optimizing a set of equalization parameters used in a DMT system. This method initializes the equalization parameters, then repeatedly sends a training sequence through the communication channel to the receiver (or decoder). Using the equalization parameters, the received sequence, and a local replica of the training sequence, the receiver updates a set of channel target response parameters. The receiver also windows the channel target response parameters as well as the equalizer parameters. The training process is repeated until a predetermined condition is met.
Although the method taught by U.S. Pat. No. 5,285,474 provides effective optimization of the equalization parameters, this method suffers from a low rate of convergence. In addition, for a training frame having 512 samples, this method requires approximately 40,000 multiplications. This requires significant computational power in practical applications. In addition, the signal-to-interference ratio (SIR) resulting from the method is not optimal.
Another method is taught by U.S. Pat. No. 5,627,863, "Frame Synchronization in Multicarrier Transmission Systems," J. Aslanis et al., May 6, 1997, which is incorporated herein by reference. Specifically, this method performs frame synchronization using correlation of frequency domain amplitudes of received synchronizing frame and stored synchronizing patterns. This method is best suited for frame synchronization during a data transmission phase, where a synchronizing frame is inserted into the data carrying frames. Using this method during the training session can require setting the time and frequency equalizer coefficients prior to the frame synchronization (see e.g., column 11, lines 41-62 of U.S. Pat. No. 5,627,863). Using an unsynchronized frame for setting the equalizer may result in a sub-optimal solution. Furthermore, this frame synchronization requires a considerable number of calculations to calculate the frame alignment.
Yet another method of generating the TEQ coefficients is taught by "Time-Domain Equalizer Training for ADSL," M. Nafie et al., IEEE International Conference on Communication, June 1997, which is incorporated herein by reference. Specifically, this method teaches off-line training techniques for generating the TEQ coefficients. The off-line training techniques include both a minimum eigenvalue approach and a time-domain LMS approach. The time-domain LMS approach requires an initial estimate of the channel impulse response. Unfortunately, under certain circumstances, this approach may generate a low-pass or zero solution for the TEQ coefficients. In addition, for a training frame having 512 samples, this approach requires approximately 30,000 multiplications. This requires significant computational power in practical applications.
Accordingly, an improved method of generating the TEQ and FEQ coefficients is desired which requires few computational resources. To this end, the method should attempt to minimize the number of comparisons required to select the correct frame alignment. In addition, the method should provide a high convergence rate. Finally, the method should provide an improved SIR.